This invention relates to a power supply circuit of a switching inverter type and, more particularly, to a power supply circuit of this type capable of reducing switching loss to the maximum and improving conversion efficiency by utilizing both voltage resonance and current resonance.
As a power supply circuit intended to reduce switching loss to the maximum and improving conversion efficiency by utilizing both voltage resonance and current resonance, there is the applicant's co-pending U.S. patent application Ser. No. 07/896,214.
Description will be made about this power supply circuit below. This power supply circuit aims at reducing switching loss to the maximum and improving conversion efficiency by utilizing both voltage resonance and current resonance as well as reducing noise by enabling operation waveforms of voltage and current appearing at respective component parts of the circuit to simulate a sine wave to a maximum degree.
This power supply circuit includes, as shown in FIG. 2, a dc power source 1, switching means 2 including switching elements which are turned on and off at desired timing, switching a dc input from the dc power source 1 to convert it to ac and providing it as an ac output from an output terminal thereof, series resonance means 4 provided in series to current flowing, from the output terminal of the switching means 2, parallel resonance means 5 provided in parallel to voltage produced at the output terminal of the switching means 2, dc output means which full-wave rectifies and smoothes an ac input supplied through the series resonance means 4 and the parallel resonance means 5 to provide a dc output, and switching control means 6 which controls the switching elements of the switching means 2 so that the switching elements turn on respectively intermittently.
FIG. 3 is a circuit diagram showing the basic construction of the power supply circuit of the invention shown in FIG. 2 somewhat more specifically. The operation of the basic construction of FIG. 3 will be described with reference to the flow chart of FIG. 4.
When switching elements S1 and S2 are being turned on and off repeatedly at a timing shown in (f) and (g) of FIG. 4, power source voltages +VI, -VI are converted into ac having substantially a peak valve of VI at a point A of the circuit of FIG. 3. This ac is rectified by diodes D1 and D2 through an inductance L2 and a capacitor C2. The rectified signal is smoothed by capacitors C3 and C4 to become dc and the dc current flows through a load RL. As the diodes D1 and D2 and the capacitors C3 and C4, the elements of substantially the same characteristics are used.
When the switching element S1 is in the on state, the diode D1 is in a forward direction so that charge current iD1 flows to the capacitor C3. Since a relation that the capacitor C3 is much larger than the capacitor C2 (C3&gt;&gt;C2) and impedances of the switching element S1 and the diode D1 are sufficiently small is established, this current iD1 becomes series resonance current of a sine waveform due to the inductance L2 and the capacitor C2 (see (b) of FIG. 4). Since reverse voltage is applied to the diode D1 and the diode D1 is turned off when the direction of reverse current is reversed upon lapse of half cycle, series resonance is stopped. In other words, when the resonance current has finished its half cycle and has become zero, the resonance is automatically stopped.
At this time, charge corresponding to the resonance current which has flown to this time point has been accumulated in the capacitor C2 and voltage VC2 across the capacitor C2 remains (see (e) of FIG. 4). Since this charge QC2=C2.multidot.VC2 is discharged to the load during a next cycle during which the switching element S2 is on state and, therefore, there is no energy loss. Since energy stored in inductance is proportional to current flowing through the inductances, energy stored in the inductance L2 is zero when the resonance has stopped at the current zero state. This signifies that generation of harmfull noise in this circuit portion is very small and also signifies an important condition under which a voltage resonance mode is established.
In order to reduce the magnetic energy of the inductance L2 completely to zero, it is necessary to keep the switching element S1 in the on state until the resonance current has become zero. It will be no use to keep the switching element S1 in the on state after the resonance current has become zero but it will be inefficient to simply prolong a time period during which energy is not transmitted. The switching element S1 therefore may be turned off with some allowance of time (TQ1-Ti). Since the resonance time (a cycle of resonance current) by the inductance L2 and the capacitor C2 is constant, a time period during which the switching element S1 is on state may also be a constant value.
When the switching element S1 is turned off, the current resonance has already finished and the current becomes zero and, therefore, current flowing through the switching element S1 at this time is only current flowing to the inductance L1. The value of the inductance L1 can be set independently from the values of the inductance L2 and the capacitor C2 and, by establishing a relation that inductance L1 is much larger than inductance L2 (L1&gt;&gt;L2), current flowing through the inductance L1 can be of a value sufficiently smaller than the resonance current of the inductance L2 and the capacitor C2 and, therefore, the switching element S1 is turned off in a nearly zero current state and, as a result, loss in turning off of the switching element S1 is extremely small. When the switching element S1 is turned off (since the switching element S2 has not been turned on, both switching elements S1 and S2 are off), the diodes D1 and D2 are also off and, therefore, the elements which are in electric operation at this time are only the inductance L1 and the capacitor C1.
Magnetic energy (current) stored in the inductance L1 during the on state of the switching element S1 constitutes energy which operates parallel resonance with the capacitor C1 which causes voltage at the point A to drop in a sine wave form and approach the potential -VI exceeding the zero point. The operation performed during this time period is the voltage resonance mode. The voltage resonance waveform in principle assumes a form as shown in (a) of FIG. 4 which is a vertically symmetrical form with respect to a point crossing the reference potential (i.e., the potential designated by "0" in (a) of FIG. 4). Depending, however, upon an actual circuit design, deformation in the waveform will take place (e.g., a case where a timing control circuit to be described later consumes a part of energy of voltage resonance through its windings).
When the potential at the point A has dropped to nearly -VI (i.e., below the potential at one end of capacitor C4), the diode D2 is turned on and thereby causes energy charged in the capacitor C4 to discharge to inductance L1 through the diode D2, the capacitor C2 and the inductance L2. Since the current flowing through the inductance L1 is set a small value, a large change in the current value does not take place but the potential at the point A remains at a value near -VI. If the switching elements S1 and S2 are kept in the off state, the magnetic energy (current) of the inductance L1 will be reduced to zero in a time length which is about half of the time period during which the switching element S1 is on and voltage across the inductance L1 (or capacitor C1) drops from the potential near -VI to zero. Alternatively stated, since the point A can be held at the potential near -VI by the magnetic energy of the inductance L1 during about half of the time period during which the switching element S1 is on, if the switching element S2 is turned on within this time period, it can perform a switching-on operation in a state where voltage across the switching element S2 is extremely small. Loss during the switching-on of the switching element S2 therefore is extremely small.
The voltage across the switching element S2 when it is turned on (i.e., difference between the above described potential near -VI and -VI) is exactly not zero but there exists voltage due to Vc2 (which is voltage across the capacitor C2) left after the current resonance during the on state of the switching element S1. The voltage Vc2 becomes a different value depending upon the value of the capacitor C2. Since the value of C2 can be set with a relatively large degree of freedom in relation to the inductance L2 and loss is generally smaller when the value of C2 is large and the value of L2 is small within a range in which series resonance can take place, the value of voltage VC2 consequently becomes small which can be almost neglected as compared with the voltage VI.
Upon turning on of the switching element S2, current resonance on the negative side takes place with a result that charge current flows to the capacitor C4. Subsequently, as shown in FIG. 3, the above operation is repeated with the switching elements S1 and S2 taking their place in turn.
Time between turning off of the switching element S1 and turning on of the switching element S2 may be set at a slightly longer time length than a time period during which the point A reaches the potential near -VI due to the voltage resonance by the inductance L1 and capacitor C1 after turning off of the switching element S1. It will be inefficient as well to take a longer time. This time period need not be set so strictly but a certain fixed value may be used.
Some more consideration will be given for the time period during which the switching elements S1 or S2 is on and the time period from turning off of the switching element S1 or S2 till turning on of the switching element S2 or S1. It may be basically said that the on period of each switching element should be set to a longer period than half cycle of resonance of the series resonance means and the off period of both switching elements should be set to a shorter period than half of the resonance period of the parallel resonance means. In this case, consideration should however be given to the amount of energy which is previously given to the voltage resonance circuit of the inductance L1 and the capacitor C1 before starting the voltage resonance mode. Consideration should also be given to the manner of determining values of the inductance L1 and the capacitor C1 even when the parallel resonance frequency is set at a certain value. The on period of each switching element determines the amount of the given energy and the off-period of both switching elements should be limited to determine by the given energy (i.e., the value corresponding to the on period). According to analysis made by the inventor, it has been found that, in actuality, once the on period and the off period have been determined, the switching frequency is determined at this time and the parallel resonance (voltage resonance) frequency satisfying the operation of this invention and a usable portion of the parallel resonance waveform are solely determined. When, for example, the on period is set to infinite small (substantial zero), the voltage resonance waveform in this case seems to change substantially in the shape of a sine waveform with substantially the same frequency as the switching frequency. It should be noted, however, that, in some cases, a desired output terminal voltage change of 2 VI is not realized notwithstanding that the voltage resonance has reached its peak value.
Further, as will be apparent from the above description, it is desirable that there should be the relations L1&gt;&gt;L2 and C2&gt;&gt;C1 as condition for setting the values of the respective resonance circuits. The rectification should be made by the full-wave rectification system. As the smoothing system, a capacitor input system should be used for the current resonance. The capacity of the smoothing capacitors C3 and C4 should be much larger than the capacitor used in the series resonance means 4 so as to prevent decrease in Q of the current resonance.
In realizing the above described basic construction as a specific circuit, as will be apparent from the above described principle, the relations L1&gt;&gt;L2 and C2&gt;&gt;C1 are desirable as actual condition for setting values of the respective resonance circuits. For satisfying these relations, primary self inductance of the transformer is effectively used as L1, an independent inductance or leakage inductance between the primary and the secondary of the transformer is used as L2. Since the rectifying circuit is positioned on the secondary side of the transformer, either a center tap type rectifying circuit or a bridge type one may be used. In any case, it must be a full-wave rectifying circuit because the current resonance must be performed with both positive and negative currents. As the smoothing circuit, a capacitor input type circuit is used for the current resonance and the relation C3&gt;&gt;C2 is maintained so as to prevent drop of Q (quality) factor in the current resonance.
The transformer viewed from the primary side is shown in FIG. 5. Since the transformer has self inductance and leakage inductance, these inductances may be utilized as L1 and L2 of FIG. 2 by properly setting the values of these inductances in the circuit design. In a general transformer, the relation L1&gt;L2 is satisfied.
The basic circuit of FIG. 3 can be modified in the form shown in FIG. 6. In FIG. 6, the current resonance is performed by the inductance L2 and the capacitor C2 which is divided in two capacitors whereas the voltage resonance is performed by the inductance L1 and the capacitor C1 which is divided in two capacitors and the inductance L1. The circuit of FIG. 5 may seem to be different from the circuit of FIG. 2 in that the loop of the voltage resonance includes both the inductance L2 and the capacitor C2 but since there are the relations L2&lt;&lt;L1 and C2&gt;&gt;C1, the presence of L2 and C2 does not substantially affect the voltage resonance and, accordingly, the voltage resonance is virtually performed by the capacitor C1 and the inductance L1 in the same manner as in the circuit of FIG. 3.
FIG. 7 shows a more specific circuit using a transformer T1 having self inductance L1 and leakage inductance L2. In this circuit, a center tap type output circuit is used as the output circuit. The center tap winding is adopted because the number of diodes in the rectifying path in each rectifying cycle thereby can be reduced and, as a result, loss due to these diodes can be held to the minimum and efficiency of the circuit as a whole can be improved. Further, two transistors are utilized as the switching elements S1 and S2. Each transitor is switched by a driving circuit having a fixed switching timing as shown in (f) and (g) of FIG. 4. Accordingly, a power supply circuit with low noise and high efficiency can be realized by a simple construction.
Benefits derived from the above described power supply circuit are summarized as follows:
First, as an advantageous result of the current resonance, noise due to current is reduced. The current noise is particularly produced when a large amount and an abrupt change in current takes place in a portion where a large current flows. The current resonance automatically stops when the current which changes in the shape of a sine wave has been reduced to zero and, therefore, very little noise is generated. As to the improvement of efficiency, the switching elements S1 and S2 are turned off at the current zero state and voltage through the diodes D1 and D2 is also reversed after the current has been reduced to zero, so that the adverse effect to the efficiency during the recovery time is reduced.
Advantageous results of the voltage resonance also are reduction of noise and improvement of efficiency of the circuit as a whole. Since parts such as semiconductors used in the power supply circuit are mounted on a chassis through an insulating material for heat radiation, electrostatic capacity may be formed by electrode-s of the parts and the chassis. The part electrode is provided with an ac signal and, accordingly, current flows to this capacity which becomes a main cause of a common mode noise. The semiconductors have also a junction capacity by itself and the inductances and transformer also have line capacity. These capacities do not appear in the circuit diagram but actually exist in the respective parts and the circuit board and, therefore, current flows to all these capacities when the circuit is in operation. Since this current is one flowing to the capacity, it becomes larger when change in the voltage (dV/dT, i.e., ratio of change in voltage V to time T) is larger. When switching is made with a square wave, this current becomes a pulse-like current generates a current noise and current flowing to the chassis causes a pulse-like common mode noise. Since this pulse-like current is supplied by the switching transistors, loss in the switching transistors is produced with resulting decrease in the efficiency. Besides, since voltage having a large dV/dT contains a high frequency component, a radio wave radiated directly from the circuit (unnecessary radiation) also becomes large.
By using the voltage resonance producing a waveform approximating a sine wave and thereby reducing dV/dT, these problems can be overcome. According to the invention, the voltage resonance is performed by the inductance L1 and the capacitor C1 only when both the switching elements S1 and S2 are on and, therefore, loss in the switching elements S1 and S2 is not produced. The current flowing through the inductance L1 and the capacitor C1 is mere transfer of mutual energy so that only reactive power is consumed and loss due to the voltage resonance is very small (theoretically zero).
For reducing a voltage type noise, it is important that dV/dT of voltage waveforms at all terminals of the circuit is neglibly small. If there is a square wave at one terminal only, it will become a noise source. Power supply circuits of a conventional voltage resonance type mostly contain square waves (e.g., in circuit portions other than a transformer output), though they have a sine wave at one spot in the circuit (e.g., in the transformer output). It is a final object of this invention to realize a practical low-noise power supply circuit and it is a feature of the invention that all voltage waveforms are similar to the voltage resonance waveform by the inductance L1 and the capacitor C1. This is achieved by performing the voltage resonance at a different time from the current resonance. That is, after reducing currents of the switching elements S1 and S2 and the diode D2 to zero and reducing also the magnetic energy of the inductance L2 to zero by the current resonance, the voltage resonance is started and, by bringing the switching elements S1 and S2 and the diodes D1 and D2 in an off state and thereby reducing current in the inductance L2 and the capacitor C2 in the voltage resonance mode to zero, the waveform at the point A and the waveform at the point A' become similar to each other. The terminal voltage waveforms of the inductance L1 and the capacitor C1 thereby become similar to the terminal waveforms of the switching elements S1 and S2, inductance L2, capacitor C2, and diodes D1 and D2 and a square wave disappears from the circuit.
The above described power supply circuit actually uses two powers and therefore requires at least two switching elements with the result that this circuit requires a complex circuit design. It is therefore difficult to apply this circuit to a switching power source of a single power source used for relatively low power applications.
It is, therefore, an object of the invention to provide a power supply circuit capable of realizing turning on at a voltage zero state and turning off at a current zero state in the same manner as in the above described power supply circuit in a switching power source of a single power source thereby achieving improvement in efficiency and reduction of noise in such power supply circuit.